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Propagation of electromagnetic waves through microstrip line can be manipulated and controlled by electromagnetic band gap (EBG) structures in the ground plane, as they exhibit both pass band and stop-band in their frequency spectrum. The main attraction of EBG is that it can simultaneously act as filter and can enhance antenna radiation properties significantly for a certain frequency band. Propagation of electromagnetic waves through a microstrip line loaded with 1D electromagnetic band gap (EBG) structures loaded dielectric slab is investigated. Usually EBG structures occupy a large circuit area as the number of cells used determines the band gap characteristics. EBG structures, also termed defected ground structure (DGS), can be applied to coplanar waveguides (CPW) for effective filtering of undesired frequency components. We have successfully designed a unit cell of DGS applied to an asymmetric coplanar wave guide (ACPW) to attain band rejection characteristics. Equivalent circuit parameters of the proposed ACPW DGS unit cell are extracted. Experimental results are validated by simulation. The proposed structure is more compact as only one ground plane of CPW is used and can be highly suitable for MMIC applications. Our new compact asymmetric coplanar waveguide filter results are published in IEE Electronic letters. Further research is going on to investigate the effect of loading EBG structure as a superstrate on microstrip line.
ACPW DGS unit cell and characterization:
A 50 O ACPW with slot width S=0.34 mm and signal strip width W=3 mm was fabricated on a substrate with thickness h=1.6 mm and dielectric constant er=4.7. A square lattice with dimension a=5 mm was etched on the lateral ground plane of the ACPW with gap length l=1 mm and gap width g=0.4 mm. The geometry of the proposed ACPW DGS unit cell is shown in Fig. 1a.
The lattice shape etched in the ground plane disturbs the current distribution, thus increasing the effective capacitance and inductance of the signal strip. Thus, the proposed DGS circuit can be represented by an equivalent LCR circuit. The capacitance and inductance of the equivalent circuit of ACPW DGS can be obtained from the following equations.
For a parallel LCR circuit at the resonant frequency:
For a parallel LCR circuit at the resonant frequency:
In these equations C is the capacitance, L the inductance, Zo characteristic impedance of the coplanar line, for the pole frequency and f2-f1 the -10 dB bandwidth of the S21 curve. The lumped element values are extracted from the simulation. The frequency response of this ACPW DGS filter obtained from IE3D simulation, experiment and extracted equivalent circuit are in good agreement and shown in Fig. 2.
Evolution of mobile communication system demands compact, low profile antennas with multiband characteristics, to meet the present day communication scenario. We have designed and successfully implemented a compact dual band folded dipole antenna suitable for DCS/PCS/2.4GHz WLAN bands. The antenna resonating at 1.87GHz and 2.46GHz has 2:1 VSWR bandwidth of 17% and 9% respectively. The proposed antenna offers 53% area reduction compared to a standard rectangular microstrip antenna. The antenna is characterized using PNA E8362B and verified numerically with simulation software Ansoft HFSS10.
The proposed microstrip-fed folded monopole antenna of dimensions l1+l2+l3, printed on FR4 substrate of relative dielectric constant εr = 4.2 and height h=1.6mm excited by a 50 Ω microstrip line of length ‘l’ with a truncated ground plane having length L and width W is shown in Fig.3.a . Monopole is placed symmetric to the ground plane (with an offset of d=11.5mm from the left end of the ground plane) as shown in Fig.3.a. For design convenience width w of the folded arm is selected as same as that of the microstrip line (In this case w=3mm). For l1=11mm l2=10.75mm, l3=7mm, L=26mm and W=8mm antenna resonates at 2.37GHz (Fig.4). But in this case, as length l1 is small impedance matching is very poor due to the coupling between the folded arm and ground plane. But by moving towards left end of the ground plane (decreasing d) impedance matching improves due to the poor coupling. In this design‘d’ is optimized as 5mm as shown in the Fig.3.b.
For achieving the second resonance another similar folded arm of dimension l4+l5+l6 is printed on the opposite side of the substrate with a separation ‘s’ as shown in Fig.3.c. The length s+ l4+l5+l6 are the resonant length corresponds to the lower frequency. In the proposed dual frequency antenna, to obtain lower resonance at 1.8GHz, dimensions are optimized as l4=11mm, l5=10.75mm, l6=12mm and s=9mm. Width w is taken as same as that of the folded arm1. For the higher resonance, folded arm2 acts as an inductive stub and hence improves the matching.
Design methodology:
From the experimental and simulation results design equations for the proposed dual frequency antenna are optimized as L=0.34λd1, W=0.12 λd1, l=0.12 λd1, l1=0.143 λd1, l2=0.14 λd1, l3=0.09 λd1, l4=0.106 λd2, l5=0.103 λd2, l6=0.115 λd2 ,d=0.065 λd1 and s=0.09 λd2. Where λd1 = λ1/√ εeff , λd2 = λ2/√ εeff, εeff = (εr+1)/2, λ1 is the free space wavelength corresponds to higher resonance (f1)and λ2 is the free space wavelength corresponding to the lower resonance(f2). For higher frequency ratios, geometrical symmetry can be maintained by adjusting the design parameters in such a way that l1+l2+l3 =0.37 λd1 and s+ l4+l5+l6 =0.41 λd2.
The measured and simulated return loss characteristics of the proposed dual frequency antenna are shown in Fig.4. Simulation was carried out using Ansoft HFSS. Antenna exhibits resonances at 1.87GHz and 2.46GHz with 2:1 VSWR bandwidths of 17.3% and 9.3% respectively. The lower resonant band extending from 1675MHz to 2000MHz (325MHz) is wide enough to cover DCS and PCS bands. The higher resonant band from 2330 MHz to 2560MHz (230 MHz) covers the 2.4GHz WLAN. Return loss characteristics of the folded monopole antenna are also shown in fig.4. At d=11.5mm monopole is resonating at 2.37GHz with poor impedance bandwidth. At d=5mm, monopole is resonating at 2.37GHz with a 2:1 VSWR bandwidth of 210MHz (2260MHz-2470MHz).
The simulated current distributions of the antenna at the two resonant frequencies are presented in Fig.6. From the figure, it is well understood that current distribution of the antenna is similar to that of a dipole at 1.8GHz. Where as at 2.4GHz, arm1 of the antenna behaves as a quarter wave monopole with arm2 as an inductive stub and hence reduces the bandwidth.
The normalized radiation patterns in the two principal planes measured at the center frequencies of the respective bands are shown in Fig.7. Antenna shows identical polarization along x direction in all the operating bands. Radiation patterns are nearly omnidirectional and approximately similar to that of an x-directed dipole. Radiation patterns at other frequencies in the respective bands are similar to those shown in Fig.7.
The propagation characteristics of a microstrip line loaded with an array of split ring resonators (SRRs) as superstrate is investigated. The presence of SRRs over the microstrip line leads to an effective negative permeability in a narrow band, where signal propagation is inhibited. The width and attenuation of the rejected frequency band depends on the height of the superstrate as well as its relative position with respect to the microstrip line.
Design details:
A 50Ω microstrip line is fabricated on a commercially available FR4 substrate characterized by the parameters εr=4.36, h=1.6mm and tanδ=0.022. SRR array of five unit cells with a period a=8mm was fabricated on Perspex substrates (εr=2.56) of thickness t=0.5mm, 1mm, 1.5mm and 2mm.The unit cell size of the structure is much smaller than the wave length of operation. Schematic diagram of a SRR unit cell having a resonance at 3.7GHz is shown in fig.9.
Evolution of mobile communication system demands compact, low profile antennas with multiband characteristics, to meet the present day communication scenario. We have designed and successfully implemented a compact dual band folded dipole antenna suitable for DCS/PCS/2.4GHz WLAN bands. The antenna resonating at 1.87GHz and 2.46GHz has 2:1 VSWR bandwidth of 17% and 9% respectively. The proposed antenna offers 53% area reduction compared to a standard rectangular microstrip antenna. The antenna is characterized using PNA E8362B and verified numerically with simulation software Ansoft HFSS10.
The proposed microstrip-fed folded monopole antenna of dimensions l1+l2+l3, printed on FR4 substrate of relative dielectric constant εr = 4.2 and height h=1.6mm excited by a 50 Ω microstrip line of length ‘l’ with a truncated ground plane having length L and width W is shown in Fig.3.a . Monopole is placed symmetric to the ground plane (with an offset of d=11.5mm from the left end of the ground plane) as shown in Fig.3.a. For design convenience width w of the folded arm is selected as same as that of the microstrip line (In this case w=3mm). For l1=11mm l2=10.75mm, l3=7mm, L=26mm and W=8mm antenna resonates at 2.37GHz (Fig.4). But in this case, as length l1 is small impedance matching is very poor due to the coupling between the folded arm and ground plane. But by moving towards left end of the ground plane (decreasing d) impedance matching improves due to the poor coupling. In this design‘d’ is optimized as 5mm as shown in the Fig.3.b.
For achieving the second resonance another similar folded arm of dimension l4+l5+l6 is printed on the opposite side of the substrate with a separation ‘s’ as shown in Fig.3.c. The length s+ l4+l5+l6 are the resonant length corresponds to the lower frequency. In the proposed dual frequency antenna, to obtain lower resonance at 1.8GHz, dimensions are optimized as l4=11mm, l5=10.75mm, l6=12mm and s=9mm. Width w is taken as same as that of the folded arm1. For the higher resonance, folded arm2 acts as an inductive stub and hence improves the matching.
Design methodology:
From the experimental and simulation results design equations for the proposed dual frequency antenna are optimized as L=0.34λd1, W=0.12 λd1, l=0.12 λd1, l1=0.143 λd1, l2=0.14 λd1, l3=0.09 λd1, l4=0.106 λd2, l5=0.103 λd2, l6=0.115 λd2 ,d=0.065 λd1 and s=0.09 λd2. Where λd1 = λ1/√ εeff , λd2 = λ2/√ εeff, εeff = (εr+1)/2, λ1 is the free space wavelength corresponds to higher resonance (f1)and λ2 is the free space wavelength corresponding to the lower resonance(f2). For higher frequency ratios, geometrical symmetry can be maintained by adjusting the design parameters in such a way that l1+l2+l3 =0.37 λd1 and s+ l4+l5+l6 =0.41 λd2.
The measured and simulated return loss characteristics of the proposed dual frequency antenna are shown in Fig.4. Simulation was carried out using Ansoft HFSS. Antenna exhibits resonances at 1.87GHz and 2.46GHz with 2:1 VSWR bandwidths of 17.3% and 9.3% respectively. The lower resonant band extending from 1675MHz to 2000MHz (325MHz) is wide enough to cover DCS and PCS bands. The higher resonant band from 2330 MHz to 2560MHz (230 MHz) covers the 2.4GHz WLAN. Return loss characteristics of the folded monopole antenna are also shown in fig.4. At d=11.5mm monopole is resonating at 2.37GHz with poor impedance bandwidth. At d=5mm, monopole is resonating at 2.37GHz with a 2:1 VSWR bandwidth of 210MHz (2260MHz-2470MHz).
It is interesting to note that both the bands are polarized along x direction. This is confirmed from s21 studies and shown in Fig.5. Cross polar level of the antenna is also very high. This will make the antenna a good choice for the proposed mobile communication applications.
The simulated current distributions of the antenna at the two resonant frequencies are presented in Fig.6. From the figure, it is well understood that current distribution of the antenna is similar to that of a dipole at 1.8GHz. Where as at 2.4GHz, arm1 of the antenna behaves as a quarter wave monopole with arm2 as an inductive stub and hence reduces the bandwidth.
The normalized radiation patterns in the two principal planes measured at the center frequencies of the respective bands are shown in Fig.7. Antenna shows identical polarization along x direction in all the operating bands. Radiation patterns are nearly omnidirectional and approximately similar to that of an x-directed dipole. Radiation patterns at other frequencies in the respective bands are similar to those shown in Fig.7.
The measured antenna gain against frequency is shown in the Fig.8. Antenna has an average gain of 2.85dBi in the DCS band, 1.94dBi in the PCS band and 1.84dBi in the WLAN band. Maximum gains observed in these bands are 4.12dBi, 2.15dBi and 2.24dBi respectively.
The propagation characteristics of a microstrip line loaded with an array of split ring resonators (SRRs) as superstrate is investigated. The presence of SRRs over the microstrip line leads to an effective negative permeability in a narrow band, where signal propagation is inhibited. The width and attenuation of the rejected frequency band depends on the height of the superstrate as well as its relative position with respect to the microstrip line.
Design details:
A 50Ω microstrip line is fabricated on a commercially available FR4 substrate characterized by the parameters εr=4.36, h=1.6mm and tanδ=0.022. SRR array of five unit cells with a period a=8mm was fabricated on Perspex substrates (εr=2.56) of thickness t=0.5mm, 1mm, 1.5mm and 2mm.The unit cell size of the structure is much smaller than the wave length of operation. Schematic diagram of a SRR unit cell having a resonance at 3.7GHz is shown in fig.9.
Experimental and simulated results:
SRR array is placed over the microstrip line as shown in fig.10(a). The transmission and reflection coefficients are measured using Agilent E8362B PNA network analyzer. The effect of lateral displacement of the SRR over the microstrip line is studied by placing the superstrate of a particular thickness at different lateral positions along the X-axis (X1,X2,X3,X4) as shown in fig.10(b). Here only one SRR element is shown for simplicity.
The observed signal inhibition in the transmission characteristics for different lateral positions of the SRR array is shown in fig. 11(a). It is found that the coupling is maximum when a larger portion of the two rings comes directly above the transmission line (X3). No stop band is observed at position X4, when SRR comes symmetrically over the microstrip line.
Experimental results with superstrates of different thickness at position X3 are shown in fig. 11(b). It is observed that the maximum stop band attenuation is obtained for minimum superstrate thickness.
From the above results, it can be inferred that the superstrate has to be placed in close proximity to the microstrip line to guarantee efficient magnetic coupling and hence maximum attenuation and band width as given in fig. 12(a). The simulated result of the structure using Ansoft HFSSTM v10 is presented in figure 12(b), which is in good agreement with the measured values.
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